FAN4803 by onsemi Datasheet | DigiKey

FAN4803 Datasheet by onsemi

FAIRCHII—D SEMICONDUCTOR w
www.fairchildsemi.com
REV. 1.2.3 11/2/04
Features
• Internally synchronized PFC and PWM in one 8-pin IC
• Patented one-pin voltage error amplifier with advanced
input current shaping technique
• Peak or average current, continuous boost, leading edge
PFC (Input Current Shaping Technology)
• High efficiency trailing-edge current mode PWM
• Low supply currents; start-up: 150µA typ., operating:
2mA typ.
• Synchronized leading and trailing edge modulation
• Reduces ripple current in the storage capacitor between
the PFC and PWM sections
• Overvoltage, UVLO, and brownout protection
• PFC V
CC
OVP with PFC Soft Start
General Description
The FAN4803 is a space-saving controller for power factor
corrected, switched mode power supplies that offers very
low start-up and operating currents.
Power Factor Correction (PFC) offers the use of smaller, lower
cost bulk capacitors, reduces power line loading and stress on
the switching FETs, and results in a power supply fully compli-
ant to IEC1000-3-2 specifications. The FAN4803 includes
circuits for the implementation of a leading edge, average
current “boost” type PFC and a trailing edge, PWM.
The FAN4803-1’s PFC and PWM operate at the same
frequency, 67kHz. The PFC frequency of the FAN4803-2 is
automatically set at half that of the 134kHz PWM. This
higher frequency allows the user to design with smaller
PWM components while maintaining the optimum operating
frequency for the PFC. An overvoltage comparator shuts
down the PFC section in the event of a sudden decrease in
load. The PFC section also includes peak current limiting for
enhanced system reliability.
Block Diagram
ISENSE
3
VEAO
4
VDC
5
ILIMIT
6
GND 2
PWM OUT 8
PFC OUT 1
–
+
–
+
COMP
COMP
35µA16.2V
17.5V
VCC
+
–
COMP
+
–
–1V
SOFT START
PFC/PWM UVLO
DUTY CYCLE
LIMIT
OSCILLATOR
PFC – 67kHz
PWM – 134kHz
VREF
VREF
1.2V
26k
40k
M1 R1 C1
30pF
M2
M7
M3
M4
M6
PWM
CONTROL
LOGIC
–
+
1.5V DC ILIMIT
PFC ILIMIT
PWM COMPARATOR
VCC OVP
PFC OFF
ONE PIN ERROR AMPLIFIER
LEADING
EDGE PFC
TRAILING
EDGE PWM
+
–
COMP
7V +
–
COMP
–1
–4
REF
VCC
7
PFC
CONTROL
LOGIC
FAN4803
8-Pin PFC and PWM Controller Combo
FAN4803 PRODUCT SPECIFICATION
2
REV. 1.2.3 11/2/04
Pin Configuration
Pin Description
1
2
3
4
8
7
6
5
PFC OUT
GND
ISENSE
VEAO
PWM OUT
VCC
ILIMIT
VDC
TOP VIEW
8-Pin SOIC (S08)
8-Pin PDIP (P08)
FAN4803
Pin Name Function
1 PFC OUT PFC driver output
2 GND Ground
3I
SENSE
Current sense input to the PFC current limit comparator
4 VEAO PFC one-pin error amplifier input
5V
DC
PWM voltage feedback input
6I
LIMIT
PWM current limit comparator input
7V
CC
Positive supply (may require an external shunt regulator)
8 PWM OUT PWM driver output
Absolute Maximum Ratings
Absolute maximum ratings are those values beyond which the device could be permanently damaged. Absolute maximum
ratings are stress ratings only and functional device operation is not implied.
Operating Conditions
Parameter Min Max Unit
I
CC
Current (average) 40 mA
V
CC
MAX 18.3 V
I
SENSE
Voltage -5 1 V
Voltage on Any Other Pin GND – 0.3 V
CC
+ 0.3 V
Peak PFC OUT Current, Source or Sink 1 A
Peak PWM OUT Current, Source or Sink 1 A
PFC OUT, PWM OUT Energy Per Cycle 1.5 µJ
Junction Temperature 150 °C
Storage Temperature Range -65°150 °C
Lead Temperature (Soldering, 10 sec) 260 °C
Thermal Resistance (
θ
JA
)
Plastic DIP 110 °C/W
Plastic SOIC 160 °C/W
Temperature Range
FAN4803CS-X 0°C to 70°C
FAN4803CP-X 0°C to 70°C
PRODUCT SPECIFICATION FAN4803
REV. 1.2.3 11/2/04
3
Electrical Characteristics
Unless otherwise specified, V
CC
= 15V, T
A
= Operating Temperature Range (Note 1)
Note:
1. Limits are guaranteed by 100% testing, sampling, or correlation with worst case test conditions.
Symbol Parameter Conditions Min TYP MAX UNITS
One-pin Error Amplifier
V
EAO
Output Current T
A
= 25°C, V
EAO
= 6V 34.0 36.5 39.0 µA
Line Regulation 10V < V
CC
< 15V, V
EAO
= 6V 0.1 0.3 µA
V
CC
OVP Comparator
Threshold Voltage 15.5 16.3 16.8 V
PFC I
LIMIT
Comparator
Threshold Voltage -0.9 -1 -1.15 V
Delay to Output 150 300 ns
DC I
LIMIT
Comparator
Threshold Voltage 1.4 1.5 1.6 V
Delay to Output 150 300 ns
Oscillator
Initial Accuracy T
A
= 25°C 60 67 74 kHz
Voltage Stability 10V < V
CC
< 15V 1 %
Temperature Stability 2 %
Total Variation Over Line and Temp 60 67 74.5 kHz
Dead Time PFC Only 0.3 0.45 0.65 µs
PFC
Minimum Duty Cycle V
EAO
> 7.0V,I
SENSE
= -0.2V 0 %
Maximum Duty Cycle V
EAO
< 4.0V,I
SENSE
= 0V 90 95 %
Output Low Impedance 8 15
Ω
Output Low Voltage I
OUT
= –100mA 0.8 1.5 V
I
OUT
= –10mA, V
CC
= 8V 0.7 1.5 V
Output High Impedance 8 15
Ω
Output High Voltage I
OUT
= 100mA, V
CC
= 15V 13.5 14.2 V
Rise/Fall Time C
L
= 1000pF 50 ns
PWM
Duty Cycle Range FAN4803-2 0-41 0-47 0-50 %
FAN4803-1 0-49.5 0-50 %
Output Low Impedance 8 15
Ω
Output Low Voltage I
OUT
= –100mA 0.8 1.5 V
I
OUT
= –10mA, V
CC
= 8V 0.7 1.5 V
Output High Impedance 8 15
Ω
Output High Voltage I
OUT
= 100mA, V
CC
= 15V 13.5 14.2 V
Rise/Fall Time C
L
= 1000pF 50 ns
Supply
V
CC
Clamp Voltage (V
CCZ
)I
CC
= 10mA 16.7 17.5 18.3 V
Start-up Current V
CC
= 11V, C
L
= 0 0.2 0.4 mA
Operating Current V
CC
= 15V, C
L
= 0 2.5 4 mA
Undervoltage Lockout Threshold 11.5 12 12.5 V
Undervoltage Lockout Hysteresis 2.4 2.9 3.4 V
FAN4803 PRODUCT SPECIFICATION
4
REV. 1.2.3 11/2/04
Functional Description
The FAN4803 consists of an average current mode boost
Power Factor Corrector (PFC) front end followed by a syn-
chronized Pulse Width Modulation (PWM) controller. It is
distinguished from earlier combo controllers by its low pin
count, innovative input current shaping technique, and very
low start-up and operating currents. The PWM section is
dedicated to peak current mode operation. It uses conven-
tional trailing-edge modulation, while the PFC uses leading-
edge modulation. This patented Leading Edge/Trailing Edge
(LETE) modulation technique helps to minimize ripple cur-
rent in the PFC DC buss capacitor.
The FAN4803 is offered in two versions. The FAN4803-1
operates both PFC and PWM sections at 67kHz, while the
FAN4803-2 operates the PWM section at twice the fre-
quency (134kHz) of the PFC. This allows the use of smaller
PWM magnetics and output filter components, while mini-
mizing switching losses in the PFC stage.
In addition to power factor correction, several protection fea-
tures have been built into the FAN4803. These include soft
start, redundant PFC over-voltage protection, peak current
limiting, duty cycle limit, and under voltage lockout
(UVLO). See Figure 12 for a typical application.
Detailed Pin Descriptions
V
EAO
This pin provides the feedback path which forces the PFC
output to regulate at the programmed value. It connects to
programming resistors tied to the PFC output voltage and is
shunted by the feedback compensation network.
I
SENSE
This pin ties to a resistor or current sense transformer which
senses the PFC input current. This signal should be negative
with respect to the IC ground. It internally feeds the pulse-
by-pulse current limit comparator and the current sense feed-
back signal. The I
LIMIT
trip level is –1V. The I
SENSE
feed-
back is internally multiplied by a gain of four and compared
against the internal programmed ramp to set the PFC duty
cycle. The intersection of the boost inductor current
downslope with the internal programming ramp determines
the boost off-time.
V
DC
This pin is typically tied to the feedback opto-collector. It is
tied to the internal 5V reference through a 26k
Ω
resistor and
to GND through a 40k
Ω
resistor.
I
LIMIT
This pin is tied to the primary side PWM current sense resis-
tor or transformer. It provides the internal pulse-by-pulse
current limit for the PWM stage (which occurs at 1.5V) and
the peak current mode feedback path for the current mode
control of the PWM stage. The current ramp is offset inter-
nally by 1.2V and then compared against the opto feedback
voltage to set the PWM duty cycle.
PFC OUT and PWM OUT
PFC OUT and PWM OUT are the high-current power driv-
ers capable of directly driving the gate of a power MOSFET
with peak currents up to ±1A. Both outputs are actively held
low when V
CC
is below the UVLO threshold level.
V
CC
V
CC
is the power input connection to the IC. The V
CC
start-
up current is 150µA . The no-load I
CC
current is 2mA. V
CC
quiescent current will include both the IC biasing currents
and the PFC and PWM output currents. Given the operating
frequency and the MOSFET gate charge (Qg), average
PFC and PWM output currents can be calculated as I
OUT
=
Qg x F. The average magnetizing current required for any
gate drive transformers must also be included. The V
CC
pin
is also assumed to be proportional to the PFC output voltage.
Internally it is tied to the V
CC
OVP comparator (16.2V)
providing redundant high-speed over-voltage protection
(OVP) of the PFC stage. V
CC
also ties internally to the
UVLO circuitry, enabling the IC at 12V and disabling it at
9.1V. V
CC
must be bypassed with a high quality ceramic
bypass capacitor placed as close as possible to the IC.
Good bypassing is critical to the proper operation of the
FAN4803.
V
CC
is typically produced by an additional winding off the
boost inductor or PFC Choke, providing a voltage that is pro-
portional to the PFC output voltage. Since the V
CC
OVP max
voltage is 16.2V, an internal shunt limits V
CC
overvoltage to
an acceptable value. An external clamp, such as shown in
Figure 1, is desirable but not necessary.
Figure 1. Optional V
CC
Clamp
V
CC
is internally clamped to 16.7V minimum, 18.3V maxi-
mum. This limits the maximum V
CC
that can be applied to
the IC while allowing a V
CC
which is high enough to trip the
V
CC
OVP. The max current through this zener is 10mA.
External series resistance is required in order to limit the
current through this Zener in the case where the V
CC
voltage
exceeds the zener clamp level.
VCC
GND
1N4148
1N4148
1N5246B
PRODUCT SPECIFICATION FAN4803
REV. 1.2.3 11/2/04
5
GND
GND is the return point for all circuits associated with
this part. Note: a high-quality, low impedance ground is
critical to the proper operation of the IC. High frequency
grounding techniques should be used.
Power Factor Correction
Power factor correction makes a nonlinear load look like a
resistive load to the AC line. For a resistor, the current drawn
from the line is in phase with, and proportional to, the line
voltage. This is defined as a unity power factor is (one). A
common class of nonlinear load is the input of a most power
supplies, which use a bridge rectifier and capacitive input fil-
ter fed from the line. Peak-charging effect, which occurs on
the input filter capacitor in such a supply, causes brief high-
amplitude pulses of current to flow from the power line,
rather than a sinusoidal current in phase with the line volt-
age. Such a supply presents a power factor to the line of less
than one (another way to state this is that it causes significant
current harmonics to appear at its input). If the input current
drawn by such a supply (or any other nonlinear load) can be
made to follow the input voltage in instantaneous amplitude,
it will appear resistive to the AC line and a unity power factor
will be achieved.
To hold the input current draw of a device drawing power
from the AC line in phase with, and proportional to, the input
voltage, a way must be found to prevent that device from
loading the line except in proportion to the instantaneous line
voltage. The PFC section of the FAN4803 uses a boost-
mode DC-DC converter to accomplish this. The input to the
converter is the full wave rectified AC line voltage. No filter-
ing is applied following the bridge rectifier, so the input
voltage to the boost converter ranges, at twice line frequency,
from zero volts to the peak value of the AC input and back to
zero. By forcing the boost converter to meet two simulta-
neous conditions, it is possible to ensure that the current that
the converter draws from the power line matches the instan-
taneous line voltage. One of these conditions is that the
output voltage of the boost converter must be set higher than
the peak value of the line voltage. A commonly used value is
385VDC, to allow for a high line of 270VACRMS. The other
condition is that the current that the converter is allowed to
draw from the line at any given instant must be proportional
to the line voltage.
Since the boost converter topology in the FAN4803 PFC is
of the current-averaging type, no slope compensation is
required.
Leading/Trailing Modulation
Conventional Pulse Width Modulation (PWM) techniques
employ trailing edge modulation in which the switch will
turn ON right after the trailing edge of the system clock.
The error amplifier output voltage is then compared with the
modulating ramp. When the modulating ramp reaches the
level of the error amplifier output voltage, the switch will be
turned OFF. When the switch is ON, the inductor current will
ramp up. The effective duty cycle of the trailing edge modu-
lation is determined during the ON time of the switch. Figure
2 shows a typical trailing edge control scheme.
Figure 2. Typical Trailing Edge Control Scheme.
RAMP
VEAO
TIME
VSW1
TIME
REF
EA
–
+
–
+
OSC
DFF
R
D
Q
Q
CLK
U1
RAMP
CLK
U4
U3
C1
RL
I4
SW2
SW1
+
DC
I1
I2 I3
VIN
L1
U2
FAN4803 PRODUCT SPECIFICATION
6REV. 1.2.3 11/2/04
In the case of leading edge modulation, the switch is turned
OFF right at the leading edge of the system clock. When the
modulating ramp reaches the level of the error amplifier
output voltage, the switch will be turned ON. The effective
duty-cycle of the leading edge modulation is determined
during the OFF time of the switch. Figure 3 shows a leading
edge control scheme.
One of the advantages of this control technique is that it
requires only one system clock. Switch 1 (SW1) turns OFF
and Switch 2 (SW2) turns ON at the same instant to mini-
mize the momentary “no-load” period, thus lowering ripple
voltage generated by the switching action. With such
synchronized switching, the ripple voltage of the first stage
is reduced. Calculation and evaluation have shown that the
120Hz component of the PFC’s output ripple voltage can be
reduced by as much as 30% using this method, substantially
reducing dissipation in the high-voltage PFC capacitor.
Typical Applications
One Pin Error Amp
The FAN4803 utilizes a one pin voltage error amplifier in the
PFC section (VEAO). The error amplifier is in reality a cur-
rent sink which forces 35µA through the output program-
ming resistor. The nominal voltage at the VEAO pin is 5V.
The VEAO voltage range is 4 to 6V. For a 11.3MΩ resistor
chain to the boost output voltage and 5V steady state at the
VEAO, the boost output voltage would be 400V.
Programming Resistor Value
Equation 1 calculates the required programming resistor
value.
PFC Voltage Loop Compensation
The voltage-loop bandwidth must be set to less than 120Hz
to limit the amount of line current harmonic distortion.
A typical crossover frequency is 30Hz. Equation 1, for
simplicity, assumes that the pole capacitor dominates
the error amplifier gain at the loop unity-gain frequency.
Equation 2 places a pole at the crossover frequency,
providing 45 degrees of phase margin. Equation 3 places a
zero one decade prior to the pole. Bode plots showing the
overall gain and phase are shown in Figures 5 and 6. Figure 4
displays a simplified model of the voltage loop.
Rp VV
I
V
AM
BOOST EAO
PGM
=−=−=
400 50V
35 113
..
µ
Ω(1)
CPin
RV VEAOC f)
COMP pBOOST OUT
=×× ××
×
×∆ (2 2
π
(2)
CW
COMP 300
11.3MΩ ×400V × 0.5V × 220µF × (2 × π × 30Hz)
F
2
16n
=
=
COMP
C
Figure 3. Typical Leading Edge Control Scheme.
REF EA
–
+
–
+
OSC
DFF
R
D
Q
Q
CLK
U1
RAMP
CLK
U4
U3
C1
RL
I4
SW2
SW1
+
DC
I1
I2 I3
VIN
L1
VEAO
CMP
U2
RAMP
VEAO
TIME
VSW1
TIME
n 4mm HHH \ .__ ~ \ \ ~
PRODUCT SPECIFICATION FAN4803
REV. 1.2.3 11/2/04 7
Internal Voltage Ramp
The internal ramp current source is programmed by way of
the VEAO pin voltage. Figure 7 displays the internal ramp
current vs. the VEAO voltage. This current source is used to
develop the internal ramp by charging the internal 30pF +12/
–10% capacitor. See Figures 10 and 11. The frequency of the
internal programming ramp is set internally to 67kHz.
PFC Current Sense Filtering
In DCM, the input current wave shaping technique used by
the FAN4803 could cause the input current to run away.
In order for this technique to be able to operate properly
under DCM, the programming ramp must meet the boost
inductor current down-slope at zero amps. Assuming the
programming ramp is zero under light load, the OFF-time
will be terminated once the inductor current reaches zero.
Rf × C
COMP COMP
1(3)
(4)
2 × π ×
RHz ×nF kΩ
COMP 1
62 8 ×30 16 330
.
CfR
ZERO
COMP
1
2 × π ××
10
CHz ×kΩµF
ZERO 1
628 ×3 330 016
..
=
==
=
=
=
Figure 4. Voltage Control Loop
FAN4803
FAN4803
IVEAO
34µA
IOUT
VO
220µF
RLOAD
667Ω330kΩ
11.3MΩ
0.15µF
15nF
POWER
STAGE COMPENSATION
VEAO
∆VEAO +
–
Figure 5. Voltage Loop Gain
60
40
20
0
–20
–40
–60
GAIN (dB)
FREQUENCY (Hz)
0.1 10 1000
1 100
Power Stage
Overall Gain
Compensation
Network Gain
Figure 6. Voltage Loop Phase
0
50
100
150
200
PHASE (º)
FREQUENCY (Hz)
0.1 1 10 1000100
Power Stage
Overall
Compensation
Network
Figure 7. Internal Ramp Current vs. VEAO
50
40
30
20
10
0
IRAMP (µA)
VEAO (V)
02 7
5
13 64
FF @ –55ºC
TYP @ –55ºC
TYP @ 155ºC
SS @ 155ºC
TYP @ ROOM TEMP
FAN4803 PRODUCT SPECIFICATION
8REV. 1.2.3 11/2/04
Subsequently the PFC gate drive is initiated, eliminating the
necessary dead time needed for the DCM mode. This forces
the output to run away until the VCC OVP shuts down the
PFC. This situation is corrected by adding an offset voltage
to the current sense signal, which forces the duty cycle to
zero at light loads. This offset prevents the PFC from operat-
ing in the DCM and forces pulse-skipping from CCM to no-
duty, avoiding DMC operation. External filtering to the cur-
rent sense signal helps to smooth out the sense signal,
expanding the operating range slightly into the DCM range,
but this should be done carefully, as this filtering also
reduces the bandwidth of the signal feeding the pulse-by-
pulse current limit signal. Figure 9 displays a typical circuit
for adding offset to ISENSE at light loads.
PFC Start-Up and Soft Start
During steady state operation VEAO draws 35µA. At start-up
the internal current mirror which sinks this current is defeated
until VCC reaches 12V. This forces the PFC error voltage to
VCC at the time that the IC is enabled. With leading edge
modulation VCC on the VEAO pin forces zero duty on the
PFC output. When selecting external compensation compo-
nents and VCC supply circuits VEAO must not be prevented
from reaching 6V prior to VCC reaching 12V in the turn-on
sequence. This will guarantee that the PFC stage will enter
soft-start. Once VCC reaches 12V the 35µA VEAO current
sink is enabled. VEAO compensation components are then
discharged by way of the 35µA current sink until the steady
state operating point is reached. See Figure 8.
PFC Soft Recovery Following VCC OVP
The FAN4803 assumes that VCC is generated from a source
that is proportional to the PFC output voltage. Once that
source reaches 16.2V the internal current sink tied to the
VEAO pin is disabled just as in the soft start turn-on
sequence. Once disabled, the VEAO pin charges HIGH by
way of the external components until the PFC duty cycle
goes to zero, disabling the PFC. The VCC OVP resets once
the VCC discharges below 16.2V, enabling the VEAO current
sink and discharging the VEAO compensation components
until the steady state operating point is reached. It should be
noted that, as shown in Figure 8, once the VEAO pin exceeds
6.5V, the internal ramp is defeated. Because of this, an exter-
nal Zener can be installed to reduce the maximum voltage to
which the VEAO pin may rise in a shutdown condition.
Clamping the VEAO pin externally to 7.4V will reduce the
time required for the VEAO pin to recover to its steady state
value.
UVLO
Once VCC reaches 12V both the PFC and PWM are enabled.
The UVLO threshold is 9.1V providing 2.9V of hysteresis.
Generating VCC
An internal clamp limits overvoltage to VCC. This clamp
circuit ensures that the VCC OVP circuitry of the FAN4803
will function properly over tolerance and temperature while
protecting the part from voltage transients. This circuit
allows the FAN4803 to deliver 15V nominal gate drive at
PWM OUT and PFC OUT, sufficient to drive low-cost
IGBTs.
It is important to limit the current through the Zener to avoid
overheating or destroying it. This can be done with a single
resistor in series with the VCC pin, returned to a bias supply
of typically 14V to 18V. The resistor value must be chosen
to meet the operating current requirement of the FAN4803
itself (4.0mA max) plus the current required by the two gate
driver outputs.
Figure 8. PFC Soft Start Figure 9. ISENSE Offset for Light Load Conditions
0
0
200ms/Div.
VBOOST
0
VOUT
VEAO
VCC 10V/div.
10V/div.
10V/div.
200V/div.
0
PFC
GATE
C23
0.01µF
CR16
1N4148
R29
20kΩ
VCC
RTN (see Figure 12)
R28
20kΩR4
1kΩ
to BR1 -Ve
C16
1µF
C5
0.0082µF
R19
10kΩ
ISENSE
R3
0.15Ω
3W
wow {PL wuuuuuuuufiwuuuuuuuuuu I: III“
PRODUCT SPECIFICATION FAN4803
REV. 1.2.3 11/2/04 9
VCC OVP
VCC is assumed to be a voltage proportional to the PFC
output voltage, typically a bootstrap winding off the boost
inductor. The VCC OVP comparator senses when this volt-
age exceeds 16V, and terminates the PFC output drive while
disabling the VEAO current sink. Once the VEAO current
sink is disabled, the VEAO voltage will charge unabated,
except for a diode clamp to VCC, reducing the PFC pulse
width. Once the VCC rail has decreased to below 16.2V the
VEAO sink will be enabled, discharging external VEAO
compensation components until the steady state voltage is
reached. Given that 15V on VCC corresponds to 400V on
the PFC output, 16V on VCC corresponds to an OVP level of
426V.
Component Reduction
Components associated with the VRMS and IRMS pins of a
typical PFC controller such as the ML4824 have been elimi-
nated. The PFC power limit and bandwidth does vary with
line voltage. Double the power can be delivered from a 220
V AC line versus a 110 V AC line. Since this is a combina-
tion PFC/PWM, the power to the load is limited by the PWM
stage.
Figure 10. Typical Peak Current Mode Waveforms
Figure 11. FAN4803 PFC Control
VISENSE
VC1 RAMP
GATE
DRIVE
OUTPUT
CZERO
ISENSE
VC1
5V
VI SENSE
GATE
OUTPUT
RCOMP
RP
VOUT = 400V
VEAO
35µAR1
4+
–
COMP
–4
3
CCOMP
C1
30pF
FAN4803 PRODUCT SPECIFICATION
10 REV. 1.2.3 11/2/04
Figure 12. Typical Application Circuit. Universal Input 240W 12V DC Output
BR1
600V
4A
GBU4J
LINE
NEUTRAL
F1 5A 250V
J1-1
J1-2
C19
4.7nF
250VAC
C20
4.7nF
250VAC
R24
470kΩ
0.5W
TH1
10Ω
5A
R3 0.15Ω 3W
102T
L2
FQP9N50
Q5
FQP9N50
Q2
FQP9N50
Q4
Q1
2N3906
1000µH
R1
36Ω
RURP860
CR1 8A, 600V
1N5818 CR7
UF4005
CR3
P6KE51CA
CR18 51V
C26
0.01µF
500V
C18 4.7nF
CR2
30A, 60V
R36 220Ω
L1 25µH
C29 0.01µF
MBR3060PT
CR2
30A
60V C2
2200µF
C3
1µF
C1
220µF
450V
R2
L3
36Ω
1N5246B
CR5
16V 0.5W
R22
10kΩ
R8 36Ω
R23
10kΩ
R38 22Ω
R30 200Ω
C7
0.1µF
R27
20kΩ
3W
R26
20kΩ
3W
R10
0.75Ω
3W
T2
T1
C23
0.01µF
CR4
UF4005
CR11
1N5818
1N5818
CR10
1N5818 CR12
CR9
1N52468
R5 36Ω
R11 150Ω
FQP9N50
Q3
R4 1kΩ
FAN4803
1
2
3
4
8
7
6
5
C15
0.015µFC6
1µF
C5
8.2nF
C28
1µF
C9
1µF
C10
2.2nF
U2
4
5
1
2
2
3
1
R17 3.3kΩ
R6 1.2kΩ
C12 0.1µF
RC431A
C25
0.01µF
500V
12V
J2-1
12VRET
J2-2
R18 1kΩ
U3
1N4148
CR8
L2
4T
1N4148
CR15
1
10
3
4
R32 100Ω
C27
0.01µF
R15
9.09kΩ
7.0V R21
10kΩ
C14
4.7µF
C17
0.1µF
R16
2.37kΩ
R13
5.8MΩ
R7
10Ω
CR16
IN4148
R12
5.8MΩ
C4
0.47µF
250VAC
R14
150Ω
2W
R37
330Ω
R9
1.5kΩ
C13 1nF
R31
10Ω
C11
1000µF
C21
1µF
C22
1µF
R29 20kΩ
R28
20kΩ
R19
10kΩ
PFC
GND
ISENSE
VEAO
PWM
VCC
ILIMIT
VDC
R20
510Ω
R25
390kΩ
C8
0.15µF
C16
0.01µF
a anagram P HHHH1 ‘ auarmsa 0223702“ or m Egg—x aasossc common" 0059 range 2%: L EM 7 gm» ‘ y—f rim, } magnum w/ WWW, W magnum itWfl a “L :
PRODUCT SPECIFICATION FAN4803
REV. 1.2.3 11/2/04 11
Mechanical Dimensions
SEA TI N G PLA N E
0.148 - 0.158
(3.76 - 4.01)
PI N 1 I D 0.228 - 0.244
(5.79 - 6.20)
0.189 - 0.19
Package: S08
8 Pin SOIC
9
(4.80 - 5.06)
0.012 - 0.020
(0.30 - 0.51)
0.050 BSC
(1.27 BSC)
0.015 - 0.035
(0.38 - 0.89)
0.059 - 0.069
(1.49 - 1.75)
0.004 - 0.010
(0.10 - 0.26)
0.055 - 0.061
(1.40 - 1.55)
8
0.006 - 0.010
(0.15 - 0.26)
0°–8°
1
0.017 - 0.027
(0.43 - 0.69)
( 4 PLA CES)
SEATING PLANE
0.240 - 0.260
(6.09 - 6.60)
PIN 1 ID 0.299 - 0.335
(7.59 - 8.50)
0.365 - 0.385
(9.27 - 9.77)
0.016 - 0.020
(0.40 - 0.51)
0.100 BSC
(2.54 BSC)
0.008 - 0.012
(0.20 - 0.31)
0.015 MIN
(0.38 MIN)
8
0° - 15°
1
0.055 - 0.065
(1.39 - 1.65)
0.170 MAX
(4.32 MAX)
0.125 MIN
(3.18 MIN)
0.020 MIN
(0.51 MIN)
(4 PLACES)
Package: P08
8-Pin PDIP
FAN4803 PRODUCT SPECIFICATION
11/2/04 0.0m 003
Stock#DS30004803
2004 Fairchild Semiconductor Corporation
LIFE SUPPORT POLICY
FAIRCHILD’S PRODUCTS ARE NOT AUTHORIZED FOR USE AS CRITICAL COMPONENTS IN LIFE SUPPORT DEVICES
OR SYSTEMS WITHOUT THE EXPRESS WRITTEN APPROVAL OF THE PRESIDENT OF FAIRCHILD SEMICONDUCTOR
CORPORATION. As used herein:
1. Life support devices or systems are devices or systems
which, (a) are intended for surgical implant into the body,
or (b) support or sustain life, and (c) whose failure to
perform when properly used in accordance with
instructions for use provided in the labeling, can be
reasonably expected to result in a significant injury of the
user.
2. A critical component in any component of a life support
device or system whose failure to perform can be
reasonably expected to cause the failure of the life support
device or system, or to affect its safety or effectiveness.
www.fairchildsemi.com
DISCLAIMER
FAIRCHILD SEMICONDUCTOR RESERVES THE RIGHT TO MAKE CHANGES WITHOUT FURTHER NOTICE TO
ANY PRODUCTS HEREIN TO IMPROVE RELIABILITY, FUNCTION OR DESIGN. FAIRCHILD DOES NOT ASSUME
ANY LIABILITY ARISING OUT OF THE APPLICATION OR USE OF ANY PRODUCT OR CIRCUIT DESCRIBED HEREIN;
NEITHER DOES IT CONVEY ANY LICENSE UNDER ITS PATENT RIGHTS, NOR THE RIGHTS OF OTHERS.
Ordering Information
Part Number PFC/PWM Frequency Temperature Range Package
FAN4803CS-1 67kHz / 67kHz 0°C to 70°C 8-Pin SOIC (S08)
FAN4803CS-2 67kHz / 134kHz 0°C to 70°C 8-Pin SOIC (S08)
FAN4803CP-1 67kHz / 67kHz 0°C to 70°C 8-Pin PDIP (P08)
FAN4803CP-2 67kHz / 134kHz 0°C to 70°C 8-Pin PDIP (P08)